Amplifier pre-distortion systems and methods

ABSTRACT

A method of optimizing performance of a multiple path amplifier includes: splitting an input signal to derive a respective sub-signal for each branch of the multiple path amplifier; independently pre-distorting each sub-signal using a known performance characteristic of its associated branch of the multiple path amplifier; and supplying each pre-distorted sub-signal to its associated branch of the multiple amplifier.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. patent application Ser. No.12/058,027 filed Mar. 28, 2008, which is claims the benefit of U.S.Provisional Patent Application No. 60/909,168 filed Mar. 30, 2007, theentire contents of which are hereby incorporated herein by reference.

TECHNICAL FIELD

The present invention relates to pre-distorting amplifiers, morespecifically to pre-distorting multiple path power amplifierarrangements.

BACKGROUND OF THE INVENTION

The voltage output of an amplifier as a function of its input voltage istypically not linear over certain ranges of input voltages, particularlylarger input voltages. Pre-distortion is a technique by which an inputsignal is pre-distorted in order to compensate for these non-linearranges of an amplifier.

One of the goals of power amplifier design for wireless base-stations isincreased efficiency. Improvements in efficiency can lead to a reducedamplifier cost (e.g. by allowing for the use of cheaper transistors withreduced power handling capability) and reduced operating expense (e.g.reduced size, reduced cooling requirements, reduced power requirements,etc.). In a conventional power amplifier, various techniques are oftenapplied to class AB output stage configurations to achieve a desiredlevel of performance, but the benefit of these techniques are limited inefficiency by the class AB output stage.

A multiple-path output stage, such as, for example, a so-called DohertyAmplifier, offers the potential of increased efficiency, but it isdifficult to pre-distort the input so as to meet demanding wirelessspecifications of such an amplifier arrangement. Further efficiencyenhancement can be achieved with an asymmetric Doherty amplifier (e.g. aDoherty amplifier arrangement where different technologies are used forthe main and peaking amplifiers) but such an arrangement furtherincreases the pre-distortion challenge.

Techniques of optimizing performance of a multiple path amplifier thatovercome at least some of the above-noted deficiencies are highlydesirable.

SUMMARY OF THE INVENTION

The present invention addresses the above-noted problems by providingtechniques for optimizing performance of a multiple path amplifier.

Thus, an aspect of the present invention provides a method of optimizingperformance of a Doherty amplifier, the method comprises: splitting aninput signal to derive a respective sub-signal for each branch of theDoherty amplifier; independently pre-distorting each sub-signal using aknown performance characteristic of its associated branch of the Dohertyamplifier; and supplying each pre-distorted sub-signal to its associatedbranch of the Doherty amplifier.

Embodiments of the present invention employ a pre-distortion mechanismthat may use knowledge of the input signal (e.g. such as envelope,amplitude, or phase, or some combination of parameters) as well as otheravailable system waveforms and characteristics (e.g. TDD state).Embodiments of the invention may provide any one or more of thefollowing: (a) a multi-input pre-distortion mechanism capable ofproducing pre-distortion characteristics, (b) the ability to createmultiple distinct pre-distorted outputs in order to support multi-inputamplifier arrangements, (c) a dynamically reconfigurable pre-distortionarchitecture that can support multiple amplifier architectures ormultiple characteristics for a single amplifier (e.g. under differentoperating conditions) (d) hardware efficient training algorithms andmechanisms that determine the optimum pre-distortion coefficients, and(e) the use of the input envelope and input signals to divide thepre-distortion into distinct regions. These regions may or may notoverlap. In this way, one may obtain improved pre-distortion performancefor multiple path power amplifier arrangements.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features and advantages of the present invention will becomeapparent from the following detailed description, taken in combinationwith the appended drawings, in which:

FIG. 1 is a block diagram schematically illustrating a typical Dohertyamplifier system known in the Prior Art;

FIG. 2 is a block diagram schematically illustrating a multiple-pathamplifier system in accordance with a first representative embodiment ofthe present invention;

FIG. 3 schematically illustrates operation of a splitter usable in themultiple-path amplifier of FIG. 2;

FIG. 4 is a block diagram schematically illustrating a multiple pathamplifier system in accordance with a second representative embodimentof the present invention;

FIG. 5 is a block diagram schematically illustrating a Three-Branchpower amplifier usable in the system of FIG. 4;

FIG. 6 is a block diagram schematically illustrating an alternativeThree-Branch power amplifier usable in the system of FIG. 4;

FIG. 7 is a block diagram schematically illustrating a generalizedN-Branch power amplifier usable in the system of FIG. 4;

FIG. 8A illustrates two types of 90° hybrid known in the art; and

FIGS. 8B-C, are block diagrams schematically illustrating possiblecombiner networks constructed using the 90° hybrids shown in FIG. 8A.

It will be noted that throughout the appended drawings, like featuresare identified by like reference numerals.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention provides pre-distortion techniques for variousconfigurations of multiple path power amplifiers. Embodiments of thepresent invention are described below, by way of example only, withreference to FIGS. 1-7.

One conventional technique applied to pre-distort a power amplifier (PA)is to treat the PA as a single input-single output block, andpre-distort the input signal Si based on a comparison of the originalsignal and the signal So at the output of the PA. Although this may beeffective where there is only ‘one path’ in the PA, it does not provideany independent correction to the multiple paths in a multiple pathpower amplifier.

Referring to FIG. 1, a classical Doherty amplifier 2 includes a signalsplitter 4; a peaking amplifier 6, a main amplifier B, and a combinernetwork 10. In operation a Radio Frequency (RF) input signal Si is splitby the analog power splitter 4 and fed to the two amplifiers 6 and 8.The respective amplified signals are then recombined by the combinernetwork 10 to produce an output signal So. Typically, the amplifierinput signal Si is generated by a linearizer 12, which is normallycascaded with an analog up-conversion block 14. As is known in the art,the linearizer 12 pre-distorts a baseband input signal Sb in order tocompensate non-linearity of the up-converter 14 and amplifier 2. Theup-conversion block 14 operates to up-convert the pre-distorted basebandsignal to radio frequency (RF).

In some cases analog linearization may be implemented, using well knowncircuit techniques. However, a more versatile arrangement implements adigital function in the linearizer 12, using coefficients that arecalculated based on a feedback signal Sf derived from the amplifieroutput signal So. The pre-distorted signal Sb′ output by the linearizer12 is then converted into an analog baseband signal by a digital toanalog converter (DAC) 16. Digital pre-distortion has the advantage thatthe coefficients can be adaptively computed, using known techniques, tooptimize the output signal So to a far greater degree of precision thanis possible using analog linearization techniques. This trainingoperation may either be performed once (e.g. during system layout andtest) or periodically during run-time, as desired. As is known in theart, digital signal pre-distortion can compensate for memory effects,and thus appropriately adjust the amplitude and phase nonlinearity ofthe signal applied to the amplifier 2.

As is known in the art, the feedback signal Sf can combined with Sb (orSi) to compute any of a variety of known error or cost functions. Thelinearizer can then operate to compute the coefficients so as tooptimize the value of the output signal So. In some cases, this meansdriving the difference between the input signal Sb and a scaled versionof the feedback signal Sf to a local minima, while in other cases thedifference is driven to zero (or some other predetermined value). Ineither case, the feedback loop may be implemented in the digital domainby either down converting the RF output signal So to baseband and thensampling the baseband signal, or by sampling the RF output signal So.The former technique allows potentially higher precision but at the costof a more complex receiver whereas the later achieves a lower costreceiver at the expense of precision.

As described above, in the classical Doherty amplifier 2 the inputsignal Si is split within the amplifier 2, downstream of the Linearizer12. In view of this arrangement, the Doherty amplifier 2 is typicallyviewed as a single input/single output block, and the linearizer must bedesigned to optimize system performance, treating both the peakingamplifier 6 and the main amplifier 8 as a single amplifier block. Thisapproach to pre-distortion means that compensation of non-linearities isdependent on a close match between the peaking and main amplifiers 6 and8. For various reasons, such as manufacturing variations, perfectsymmetry between the peaking and main amplifiers 6 and 8 is verydifficult to achieve, so the degree of signal correction that can beobtain is limited. Additionally, this also limits the ability to usedifferent (not matched) amplifiers (transistors).

The present invention overcomes these issues by providing systems inwhich each path of a multiple path amplifier are independentlypre-distorted. Representative embodiments of the invention are describedbelow.

FIG. 2 shows a first representative embodiment of a system in accordancewith the present invention. As may be seen in FIG. 2, a modified Dohertyamplifier 18 is similar to the classical Doherty Amplifier 2 of FIG. 1,except that the analog splitter 4 has been removed. This means that themodified Doherty amplifier 18 is a two-input/single output device, inwhich both branches 6,8 are available for independent pre-distortion. Acorresponding digital linearizer 20 is provided, which includes adigital signal processing function (DSP) 22 for processing a basebandinput signal Sb to derive the respective branch sub-signals Sp and Sm,each of which is then processed by a respective pre-distortion block 24.Each pre-distortion block 24 implements a digital linearization functionbased on respective coefficients calculated using a feedback signal Sf,in a manner similar to that described above. In this case, however, thecoefficients supplied to each pre-distortion block 24 can be computed tooptimize each of the peaking and main amplifier paths independently ofone another. A respective DAC 18 (not shown in FIG. 2) and analogup-converter 14 can be provided for up-converting each pre-distortedbranch sub-signal to RF upstream of the modified Doherty amplifier 18.

The digital signal processing function 22 may implement any of a varietyof techniques for deriving the branch sub-signals Sp and Sm from thebaseband input signal Sb. For example, the DSP 22 may implement anamplitude splitting scheme using a predetermined threshold, as will bedescribed below in greater detail. Alternatively, a power splittingscheme may be implemented, such that the power levels of the two branchsub-signals are equal (or follow some other desired relationship). Othermathematical functions could also be used to generate the branchsub-signals, without departing from the present invention.

As may be appreciated, by moving the signal splitting function into thedigital domain at the linearizer 20, a much more complex andcontrollable split can be achieved, as compared to an analog powersplitter. One benefit is that the respective sub-signals Si(p) and Si(m)supplied to the peaking and main amplifiers 6 and 8 can be pre-distortedwith different sets of linearizer coefficients, to allow much betterlinearization to be achieved. This allows flexibility in the design andselection of the main and peaking amplifiers 6 and 8, so that, forexample, symmetry between the two amplifiers is no longer required.

In the embodiment of FIG. 2, the feedback signal Sf is derived from theoutput signal So, in a manner directly analogous to that of FIG. 1. Inthe embodiment of FIG. 2, however, the feedback signal Sf is used tocalculate respective different pre-distortion coefficients for eachbranch sub-signal Sp and Sm. If desired, the feedback signal Sf mayalternatively be derived from the output of each of the peaking and mainamplifiers (i.e. by tapping the output of each amplifier upstream of thecombiner 10), either alone or in combination with the output signal So.As a still further alternative, other amplifier parameters (e.g.amplifier voltage, current, gate bias etc.) may be used alone or incombination with any other parameters for this purpose.

By way of example only, FIG. 3 schematically illustrates an amplitudesplitting scheme, in which the baseband input signal Sb (shown as ananalog signal for convenience of illustration only) is divided into apeaking signal Sp and a main signal Sm using a predetermined slicingthreshold. This splitting method follows the classical Doherty design,in which the main amplifier 8 amplifies most of the of the signal,whereas the peaking amplifier 6 amplifies only signal peaks above apredetermined threshold (which classically corresponds with saturationof the main amplifier 8). With the arrangement of FIG. 2, respectivedifferent pre-distortion coefficients can be calculated for each branch,so as to optimize the performance of each branch independently of theother. This approach reduces the matching requirements for the main andpeaking amplifier designs while improving performance. In addition, theuse of a DSP 22 means that the slicing threshold can be adaptivelyadjusted, as desired, to aid in optimization of the overall systemperformance.

As may be seen in FIG. 3, each of the branch sub-signals exhibit sharptransitions (discontinuities) at the slicing threshold point. In someembodiments, the pre-distortion (or linearization) function implementedby the pre-distorters 24 may modify these transitions to minimizespurious emissions (noise) created by the abrupt transitions, whileensuring that the final combined output signal faithfully represents theinput signal scaled by the overall amplifier gain.

FIGS. 4 and 5 show a representative transmitter system utilizing athree-branch amplifier 26, in which three amplified sub-signals arecombined to generate two output signals. In this case, the DSP 22 of thelinearizer 20 is designed to process two input signals (Sb1 and Sb2) toderive three branch sub-signals, which are then pre-distorted asdescribed above before being amplified and combined into twocorresponding output signals (So1 and So2). In a WiMAX system, forexample, the input signals Sb1 and Sb2 may respectively represent mainand diversity signals. As will be appreciated, in this case, the use ofDSP 22 in the linearizer facilitates implementation of a complexmathematical relationship between the two input signals Sb1 and Sb2 andthe branch sub-signals. For example, the DSP 22 may operate to processthe two input signals Sb1 and Sb2 to generate respective main signals Aand B, and a common peaking signal C.

Also shown in FIG. 4 is multi-path Digital Up-conversion, RF DACs andFilters (e.g. Surface Acoustic Wave (SAW) filters), all of which operatein a conventional manner, and do not require detailed descriptionherein.

As may be seen in FIG. 5, the three-branch amplifier 26 receives thethree analog signals (A, B and C) from the linearizer 20; amplifies themusing respective amplifiers 28, and then combines the amplified signalsusing a combiner network 30, to generate the desired output signals So1and So2. As will be appreciated, the design of the combiner network willbe designed jointly with the signal processing function implemented bythe DSP 22 to derive the three branch signals A B and C.

Finally, each of the output signals So1 and SO2 can be used to deriverespective feedback signals Sf1 and Sf2 for each pre-distorter 24 of thelinearizer 20, as shown in FIG. 5. If desired, feedback for thepre-distorters 24 may be based on the output signals So1 and SO2, eitheralone or in combination with internal signals such as, for example, theamplified branch sub-signals (at the output of each amplifier 28), orother internal amplifier measurements, as described above.

FIG. 6 shows another transmitter system in accordance with the presentinvention, which utilizes a three branch amplifier 26. In the embodimentof FIG. 6, a 2 input/2 output configuration is illustrated similar tothat of FIGS. 4 and 5. However, the utilization of the splitting schemeimplemented by the DSP 22 is flexible and can be chosen to reflect thedesired amplifier arrangement. For example, one peaking amplifier plus 2main amplifiers (as described above with reference to FIGS. 4 and 5),and two peaking amplifiers plus one main amplifier, or three similaramplifiers where the linearization algorithm minimizes the usage of eachin some beneficial way are all possible arrangements. Thus, the branchsub-signals A, B, and C can be defined as respective functions of thetwo inputs: e.g. A=f1(Sb1, Sb2), B=f2(Sb1, Sb2), and C=f3(Sb1, Sb2).During run-time, the DSP 22 operates to process the two baseband inputsignals (Sb1, Sb2) in accordance with the defined functions f1, f2, f3to generate the three branch sub-signals (A, B, C) Similarly, the outputsignals So1-G*Sb1 and So2-G*Sb2 can be defined as functions of theamplified branch sub-signals g1*A, g2*B and g3*C (for example,So1=fm(g1*A, g2*B, g3*C) and So2=fd(g1*A, g2*B, g3*C)), which areimplemented by the combining network 30 of the multiple path amplifier26.

As may be appreciated, by splitting/combining signals in this manner, atransmitter can be realized with fewer parallel branches than would berequired if the classical signal splitting arrangement (i.e. respectivemain and peaking paths for each input signal) of FIGS. 1-3 was used.

FIG. 7 shows a generalized version of the transmitter of FIG. 6, wherethere are N input signals Sb1 . . . SbN, which are processed by thetransmitter to yield N corresponding output signals So1 . . . SoN. TheDSP 22 implements a digital splitting function to generate M branchsub-signals S1 . . . SM, each of which is a predetermined function ofthe N input signals. The combiner network 30 at the amplifier outputoperates to combine the M amplified branch sub-signals to produce thedesired N output signals. As will be appreciated, the splitting functionimplemented by the DSP 22 is related to the desired amplifierarrangement, and the combining network 30 is matched to the splittingfunction so that the output signals So1 . . . SoN properly correspondwith the input signals Sb1 . . . SbN.

For this generalized case, the number of branch sub-signals can rangefrom:

For N=1: M=2

For N>=2: M>=N

The case of M=N+1 corresponds, for example, to an amplifier arrangementin which N amplifiers are used to amplify a respective main signalcomponent Sm of each baseband input signal, while a common peakingamplifier is used to amplify a composite peaking signal which is the sumof the peaking signal components of all of the input signals.

In the foregoing description, the equalizer 20 is described asimplementing a two-step process, in which the process of generating theindividual branch signals and then pre-distorting the branch signals areperformed as two sequential steps. However, those of ordinary skill inthe art will recognise that the DSP 22 and pre-distorters 24 can beimplemented using a single physical device (such as, for example, aRandom Access Memory look-up table). In such a case, both the signalsplitting and pre-distortion steps can be mathematically merged into asingle operation.

A further alternative may be to conceptually reverse the order ofoperations. In such a case, the respective different portions of theinput signal can be defined (for example based on a thresholdcomparison, as illustrated in FIG. 3), such that each defined portioncorresponds with a respective one of the sub-signals. The Input signalcan then be processed to pre-distort each portion of the signal, basedon the performance characteristics of its associated branch of themultiple path amplifier. Finally, each of the predistorted portions ofthe input signal can then be supplied to the respective branches of themultiple path amplifier.

As described above, the digital equalizer 20 is capable of applyingvirtually any desired mathematical operation to the input signal(s) So.As such, the DSP 22 can implement any desired mathematical function forgenerating the branch signals. The classical “amplitude splitting”function described above with reference to FIG. 3 is one such function,but the DSP 22 is not limited to this. As will be appreciated, inpractice, the primary limitations on the functions used to generate thebranch sub-signal, are imposed by the signal combiner 30, which is ananalog signal combiner network. Thus, which it is possible to define anynumber of mathematical functions by which N input signals Sb may betransformed into M sub-signals, which are amplified and thensubsequently recombined into the desired output signals So, only some ofthese relations will be realizable in a practical analog combinernetwork. Accordingly, it is anticipated that the combiner network 30will be designed first, in order to obtain a practically realizablesignal combining function, and then the complementary (signal splitting)function implemented by the DSP 22 will be designed.

For example, FIG. 8A shows two types of conventional 90° hybrid, wellknown in the art, namely: a 90° hybrid 32 and a 4.78 dB 90° hybrid 33.As is well known in the art, each hybrid receives two inputs (labelledas “in” and “Isol.” in FIG. 8A), and produces two outputs (labelled asSo1 and So2 in FIG. 8A). For the 90° hybrid 32,

${{{So}\; 1} = {{\frac{1}{\sqrt{2}}{in}} - {j\frac{1}{\sqrt{2}}{Isol}}}},{and}$${{So}\; 2} = {{\frac{1}{\sqrt{2}}{Isol}} - {j\frac{1}{\sqrt{2}}{{in}.}}}$

For the 4.78 dB 90° hybrid 33,

${{{So}\; 1} = {{\frac{1}{\sqrt{3}}{in}} - {j\sqrt{\frac{2}{3}}{Isol}}}},{and}$${{So}\; 2} = {{\frac{1}{\sqrt{3}}{Isol}} - {j\sqrt{\frac{2}{3}}{{in}.}}}$

As shown in FIG. 8B, the 90° hybrids 32 and 33 can be interconnected,along with phase delays 34, to form a combiner network 30 thatimplements the function:

$\begin{bmatrix}{{So}\; 1} \\{{So}\; 2} \\{{So}\; 3}\end{bmatrix} = {{\frac{1}{\sqrt{3}}\begin{bmatrix}{{- 60}{^\circ}} & {{- 90}{^\circ}} & {{- 180}{^\circ}} \\{{- 150}{^\circ}} & {60{^\circ}} & {{- 150}{^\circ}} \\{120{^\circ}} & {{- 150}{^\circ}} & {{- 120}{^\circ}}\end{bmatrix}}\begin{bmatrix}A \\C \\B\end{bmatrix}}$

For embodiments in which three branch sub-signals are to be combined toyield two output signals, then one of the output signals produced by thecombiner network (e.g. So3) corresponds to a “don't care” condition, andso can be terminated. The complementary function, for splitting twobaseband input signals (e.g. Sb1 and Sb2) into the branch signals A, Band C can then be derived from the combiner function using knowntechniques, for implementation by the DSP 22.

If desired, the combiner network 30 of FIG. 8B, may be further combined,to yield the 6×6 combiner network of FIG. 8C, which has the combinerfunction:

$\begin{bmatrix}{{So}\; 1} \\{{So}\; 2} \\{{So}\; 3} \\{{So}\; 4} \\{{So}\; 5} \\{{So}\; 6}\end{bmatrix} = {{\frac{1}{\sqrt{6}}\begin{bmatrix}{{- 60}{^\circ}} & {{- 90}{^\circ}} & {180{^\circ}} & {{- 150}{^\circ}} & {180{^\circ}} & {90{^\circ}} \\{{- 150}{^\circ}} & {60{^\circ}} & {{- 150}{^\circ}} & {120{^\circ}} & {{- 30}{^\circ}} & {120{^\circ}} \\{120{^\circ}} & {{- 150}{^\circ}} & {{- 120}{^\circ}} & {30{^\circ}} & {120{^\circ}} & {150{^\circ}} \\{{- 150}{^\circ}} & {180{^\circ}} & {90{^\circ}} & {- {60 \circ}} & {{- 90}{^\circ}} & {180{^\circ}} \\{120{^\circ}} & {{- 30}{^\circ}} & {120{^\circ}} & {{- 150}{^\circ}} & {60{^\circ}} & {{- 150}{^\circ}} \\{30{^\circ}} & {120{^\circ}} & {150{^\circ}} & {120{^\circ}} & {{- 150}{^\circ}} & {{- 120}{^\circ}}\end{bmatrix}}\begin{bmatrix}A \\B \\C \\\begin{matrix}D \\E\end{matrix} \\F\end{bmatrix}}$

As may be seen, this combiner matrix combines M=6 branch sub signals A .. . F to yield six output signals So1 . . . So6. Here again, in anembodiment in which fewer than six output signals are desired (i.e.,N≦6), the unused output(s) are simply terminated. The complementaryfunction, for splitting N<˜6 input signals Sb into M=6 branch subsignals A . . . F can then be derived from the combiner function usingknown techniques, for implementation by the DSP 22.

As may be appreciated, the pattern of FIGS. 8B and 8C may be repeated,as desired, to construct combiner networks of larger sizes, if desired.

The embodiments of the invention described above are intended to beillustrative only. The scope of the invention is therefore intended tobe limited solely by the scope of the appended claims.

1. A method of optimizing performance of a multiple path amplifier, themethod comprising: digitally processing an input signal to generate arespective sub-signal for each branch of the multiple path amplifier;digitally pre-distorting each sub-signal using a known performancecharacteristic of its associated branch of the multiple path amplifier;and supplying each pre-distorted sub-signal to its associated branch ofthe multiple path amplifier.
 2. A method as claimed in claim 1, whereinthe steps of digitally processing the input signal to generate arespective sub-signal for each branch of the multiple path amplifier anddigitally pre-distorting each sub-signal are performed sequentially. 3.A method as claimed in claim 1, wherein the steps of digitallyprocessing the input signal to generate a respective sub-signal for eachbranch of the multiple path amplifier and digitally pre-distorting eachsub-signal are performed as a single operation.
 4. A method as claimedin claim 1, wherein the steps of digitally processing the input signalto generate a respective sub-signal for each branch of the multiple pathamplifier and digitally pre-distorting each sub-signal comprise:defining a respective portion of the input signal corresponding to eachsub-signal; pre-distorting each of the defined portions of the inputsignal, using the known performance characteristic of its associatedbranch of the multiple path amplifier; and supplying each of thepredistorted portions of the input signal to the respective branches ofthe multiple path amplifier.
 5. A method as claimed in claim 1, whereindigitally processing the input signal comprises: comparing the inputsignal to a predetermined threshold; and splitting the input signal intoa corresponding first sub-signal and a second sub-signal based on thecomparison result.
 6. A method as claimed in claim 5, further comprisingperiodically adjusting a value of the predetermined threshold.
 7. Amethod as claimed in claim 1, wherein digitally processing the inputsignal comprises splitting the input signal into a corresponding firstsub-signal and a second sub-signal such that respective power levels ofthe first sub-signal and the second sub-signal satisfy a predeterminedrelationship.
 8. A method as claimed in claim 7, wherein thepredetermined relationship is equality between the power levels of thefirst sub-signal and the second sub-signal.
 9. A method as claimed inclaim 1, wherein pre-distorting each sub-signal comprises, for eachsub-signal, digitally processing the sub-signal using a respective setof predetermined coefficients.
 10. A method as claimed in claim 9,wherein the respective set of predetermined coefficients are adaptivelycomputed to optimize at least one parameter of the associated branch ofthe multiple path amplifier.
 11. A method as claimed in claim 10,wherein the at least one parameter comprises any one or more of: adifference between the input signal and a corresponding output signal ofthe multiple path amplifier; a difference between the input signal and acorresponding amplified sub-signal within the associated branch of themultiple path amplifier; an amplifier parameter of the associated branchof the multiple path amplifier.
 12. A method as claimed in claim 11,wherein the amplifier parameter of the associated branch comprises anyone or more of an amplifier voltage, amplifier current and gate bias.13. A method as claimed in claim 10, wherein the respective set ofpredetermined coefficients are computed once.
 14. A method as claimed inclaim 10, wherein the respective set of predetermined coefficients areperiodically updated.
 15. A method as claimed in claim 1, wherein themultiple path amplifier comprises M, where M>2, branches for amplifyingM corresponding sub-signals, and a combiner network for combining the Mamplified sub-signals from each branch to generate N, where N>1, outputsignals, and wherein digitally processing the input signal comprises:defining each one of the M sub-signals as a respective predeterminedfunction of N input signals; and for each sub-signal, digitallyprocessing the N input signals using the respective predeterminedfunction.
 16. A digital linearizer for optimizing performance of amultiple path amplifier, the digital linearizer comprising: a digitalsignal processor for digitally processing an input signal to generate arespective sub-signal for each branch of the multiple path amplifier;and a respective pre-distorter for digitally pre-distorting eachsub-signal using a known performance characteristic of its associatedbranch of the multiple path amplifier, an output of each pre-distorterbeing connected to supply a pre-distorted sub-signal to its associatedbranch of the multiple path amplifier.
 17. A digital linearizer asclaimed in claim 16, wherein the digital signal processor and therespective pre-distorter for each sub-signal are provided as separatecomponents.
 18. A digital linearizer as claimed in claim 16, wherein thedigital signal processor and the respective pre-distorter for eachsub-signal are provided as a single component.
 19. A digital linearizeras claimed in claim 18, wherein single component comprises a RandomAccess memory Look-Up Table.